Photonic implementation of jamming avoidance response

ABSTRACT

Various examples are provided for jamming avoidance response (JAR), and photonic implementations thereof. In one example, a method includes generating optical pulses that correspond to raising envelope of a beat signal associated with an interference signal and a reference signal; generating optical spikes that correspond to positive zero crossing points of the reference signal; and providing a phase output that indicates whether the beat signal is leading or lagging the reference signal, the phase output based at least in part upon the optical spikes. An adjustment to a reference frequency can be determined based at least in part upon the optical pulses and the phase output. In another example, a JAR system includes photonic circuitry to generate the optical pulses; photonic circuitry to generate the optical spikes; and photonic circuitry to provide the phase output. A logic unit can determine the adjustment to the reference frequency.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of co-pending U.S. patentapplication Ser. No. 15/335,672, entitled “PHOTONIC IMPLEMENTATION OFJAMMING AVOIDANCE RESPONSE,” filed Oct. 27, 2016, which claims priorityto, and the benefit of, U.S. provisional application entitled “PhotonicImplementation of Jamming Avoidance Response” having Ser. No.62/246,903, filed Oct. 27, 2015, both of which are hereby incorporatedby reference in their entireties.

BACKGROUND

The radio frequency (RF) spectrum is a scarce resource for both licensedand unlicensed applications. The prevalence of wireless devices and theprogression of communications technology have facilitated the vision ofanytime, anywhere access to wireless networks, as well as for radarsystem to gain ubiquitous access to the target. Ubiquitous access hasrecast modern communication and introduced a new level of strain onnetworks, making such systems more susceptible to radio interference,unfriendly jamming, and advertent jamming. Traditionally, the FederalCommunications Commission (FCC) has controlled the radio frequencyspectrum in an effort to minimize these issues by allotting bands todifferent applications and users, including commercial, defense, andcivilian applications. Considering the recent boom in wirelesstechnologies, those unlicensed bands left open for public use are oftenover-crowded and consequently suffer from radio interference, which isdifficult to mitigate due to the wide range of devices that operate atsuch frequencies. Further licensing of personal devices would aid inminimizing interference; however, it would severely reduce theflexibility of the public bands. The frequency ranges open for personaland commercial use are limited, and placing additional restrictionscould prove too complex of a task that leads to further complications.

BRIEF DESCRIPTION OF THE DRAWINGS

Many aspects of the present disclosure can be better understood withreference to the following drawings. The components in the drawings arenot necessarily to scale, emphasis instead being placed upon clearlyillustrating the principles of the present disclosure. Moreover, in thedrawings, like reference numerals designate corresponding partsthroughout the several views.

FIG. 1 is a plot illustrating an example of a beat signal and areference signal, in accordance with various embodiments of the presentdisclosure.

FIG. 2A is a schematic diagram illustrating an example of Eigenmannianeural circuitry for jamming avoidance response (JAR), in accordancewith various embodiments of the present disclosure.

FIG. 2B is a table illustrating JAR decisions based on P-unit and TSoutputs of the Eigenmannia neural circuitry of FIG. 2A, in accordancewith various embodiments of the present disclosure.

FIGS. 3A and 3B illustrate an example of TS operation principle, inaccordance with various embodiments of the present disclosure.

FIG. 4 is a schematic diagram illustrating an example of photonic JARcircuitry, in accordance with various embodiments of the presentdisclosure.

FIG. 5A is a schematic diagram illustrating an example of P-unitcircuitry of the photonic JAR circuitry of FIG. 4, in accordance withvarious embodiments of the present disclosure.

FIGS. 5B through 5K illustrate operation of the P-unit of FIG. 5A, inaccordance with various embodiments of the present disclosure.

FIG. 6A is a schematic diagram illustrating an example of ELL/T-unitcircuitry of the photonic JAR system of FIG. 4, in accordance withvarious embodiments of the present disclosure.

FIGS. 6B through 6D illustrate operation of the ELL/T-unit circuitry ofFIG. 6A, in accordance with various embodiments of the presentdisclosure.

FIG. 7A is a schematic diagram illustrating an example of a TS unitcircuitry of the photonic JAR circuitry of FIG. 4, in accordance withvarious embodiments of the present disclosure.

FIGS. 7B through 7E illustrate operation of the TS unit of FIG. 7A, inaccordance with various embodiments of the present disclosure.

FIG. 8 is a table illustrating JAR decisions based on P-unit and TSinputs of the photonic JAR circuitry of FIG. 4, in accordance withvarious embodiments of the present disclosure.

FIGS. 9A through 9H illustrate examples of inputs for variousinterference scenarios of the photonic JAR circuitry of FIG. 4, inaccordance with various embodiments of the present disclosure.

FIG. 10A is a schematic diagram illustrating an example of the photonicJAR circuitry used for testing, in accordance with various embodimentsof the present disclosure.

FIGS. 10B-10F shows waveforms generated by the photonic JAR circuitry ofFIG. 10A, in accordance with various embodiments of the presentdisclosure.

FIG. 10G is a schematic diagram illustrating an example of the logic andcontrol circuitry of the photonic JAR circuitry of FIG. 10A, inaccordance with various embodiments of the present disclosure.

FIGS. 10H-10L shows waveforms generated by the photonic JAR circuitry ofFIG. 10A, in accordance with various embodiments of the presentdisclosure.

FIGS. 11A and 11B are schematic diagrams illustrating an example ofanother photonic JAR system used for testing, in accordance with variousembodiments of the present disclosure.

FIGS. 11C-11F shows waveforms generated by the photonic JAR system ofFIGS. 11A and 11B, in accordance with various embodiments of the presentdisclosure.

FIG. 11G is a table illustrating JAR decisions based on amplitude andphase relationships, in accordance with various embodiments of thepresent disclosure.

FIGS. 11H-11M shows radio frequency spectrum as a result of the photonicJAR system of FIGS. 11A and 11B, in accordance with various embodimentsof the present disclosure.

DETAILED DESCRIPTION

Disclosed herein are various embodiments related to jamming avoidanceresponse (JAR). Reference will now be made in detail to the descriptionof the embodiments as illustrated in the drawings, wherein likereference numbers indicate like parts throughout the several views.

Dynamic spectrum access, opportunistic spectrum access, spectralpartitioning, and channelizing represent proposed approaches towardstackling this issue, but all require some degree of controlcommunication between devices, which could similarly be impacted byinterference due to overcrowding. Another major research focus,cognitive radio, involves the use of weak probe signals that scan thespectrum for “spectral holes,” regions of the spectrum not beingactively used. Spectrum sensing over a large bandwidth can be difficult,however, due to difficulties arising from avoiding interference causedby the probing with an individual device over a range of frequencies. Adynamic approach to managing spectrum scarcity, as opposed to staticspectrum allocation, can allow for efficient use of the frequencyspectrum for radio frequency (RF) communications.

The approach of the present disclosure was inspired by the neuralcircuitry of the Eigenmannia, a genus of fish, and avoids thepossibility of causing interference through probing and comprehensivespectral scanning. The Eigenmannia use electrolocation to determinetheir surroundings by generating electric fields and detectingdisturbances in the field caused by nearby objects. For these creatures,their ability to effectively make sense of their surroundings is a lifeor death necessity, and their neural circuitry has evolved to reflectthis. An individual fish emits and receives a low frequency electricalsignal in the low kilohertz range, while simultaneously being able tosense the frequency output of other nearby Eigenmannia, andautomatically regulates its own frequency in an effort to avoidinterference and jamming.

This regulation of frequency, known as jamming avoidance response (JAR),represents a method of uncoordinated communication that can be appliedin modern wireless and radar systems, eliminating the need for furtherrestriction of the unlicensed FCC bands and allowing for maximumspectrum efficiency in all bands. The ability for an individual unit toadjust its output frequency based on observed interference can beachieved with no direct coordination with other units in the system, andis therefore ideal for the cluttered complexity that defines unlicensedfrequency bands. The JAR approach towards detecting and using spectralholes is a practical means of solving a physical problem that avoidsindirectly, incompletely mitigating the issue by managing networks withexcessive protocol.

A photonic approach to JAR is ideal considering the flat frequencyresponse of photonic devices in the terahertz range, theirelectromagnetic (EM) immunity, and near-instantaneous response times.The uniform frequency response is of particular importance in thisscenario considering the lack of a priori knowledge of which frequenciesare available in a system. Furthermore, processing of unknown signals inthe RF range is not efficient and often impossible electronically due tothe bandwidth limitation and precise design needed in electronics fordifferent frequency bands. On the other hand, it is possible tooptically achieve this due to the consistent performance of opticaldevice over a wide range of frequencies. In this disclosure, a photonicJAR circuit is developed through simulation and experiment using linearand nonlinear optical effects and techniques used in photonicneuromorphic processing systems. The device can serve as a primitiveexample of a solution to the spectral scarcity and inadvertent jammingissues that will continue to build in severity if not handled on thephysical level. With the photonic JAR, radio interference and jammingcan be avoided and a more spectrally efficient, automatic, andself-adaptive RF communication system can be realized.

As previously stated, the Eigenmannia genus represents a categorizationof weakly electric fish that use electrolocation as a means of sensingand moving through their surroundings in the ocean. An Eigenmannia fishalso discharge a low frequency signal to communicate with otherEigenmannia nearby. If another fish is nearby and emitting a signal at asimilar frequency, the two fish will be jamming each other due tointerference, disable their ability to communicate. Accordingly, theirneural circuitry is able to process amplitude and phase information ofthe interference signal, determine whether it's higher or lower infrequency relative to the fish's own discharge frequency, and thenautomatically shift its discharge to a different operating frequency.Interestingly, this adjustment is always made in the proper direction(i.e., the operating frequency will be lowered if the interfering signalis higher in frequency than the reference or raised if the interferingsignal is lower in frequency). This implies that the process is not oneof haphazard, random switching but a systematic response to accurateinformation drawn from the detected interference waveform.

The process by which JAR takes place is most easily described by aphasor phenomenon. When a reference sinusoid signal at frequency f_(E)interfered by another sinusoid signal of similar frequency f_(I), a beatenvelope signal results at a frequency |f_(b)|=f_(E)−f_(I). Referring toFIG. 1, shown is a plot illustrating an example of a 20 MHz beat signal(solid line 103) alongside a 100 MHz reference sinusoid signal (dashedline 106), generating from the reference beating with an 80 MHzinterfering sinusoid signal.

As illustrated by FIG. 1, the individual peaks of the 20 MHz beat 103rotate clockwise around the peaks of the 100 MHz reference signal 106 ifthe interfering signal is at a lower frequency of 80 MHz, e.g., atpoints (i)-(vi) in FIG. 1. The rotation direction uniquely determinesthe sign of the frequency difference.

FIG. 2A shows a simplified schematic diagram illustrating the neuralcircuitry of the Eigenmannia that achieves the JAR. The neural circuitryincludes P, T, E, and I-units, the electrosensory lateral line lobe(ELL) 203, the torus semicircularis (TS) 206, nucleus electrosensorius(nE) 209, pacemaker nucleus (Pn) 212, and the electric organ (EO) 215.The P, E, and I-units convey beat amplitude information and the T-unitand components of the TS 206 convey phase information. Logic performedby other TS elements excite the nE+ or nE− neurons 209, increasing ordecreasing the frequency of the discharge regulated by the Pn 212, andultimately transmitted by the EO 215.

The simplified Eigenmannia JAR neural circuitry begins with twodifferent types of electroreceptors located on the body's surface, theT-unit and the P-unit. The P-unit is primarily responsible forinterpreting information about the amplitude of beat signal and consistsof a cluster of neurons that fire rapidly when the beat is increasing inamplitude and fire slowly when the beat is decreasing in amplitude. TheT-unit neurons process phase information of the reference signal byfiring at every positive zero crossing point of reference discharge. Theoutputs of the T-unit and P-unit receptors are both sent to the ELL 203,in which the amplitude and phase information are independentlyprocessed. As shown in FIG. 2A, the ELL 203 includes the E and I-units.A rapid firing rate of the P-unit results in the E-unit firing and aslow firing rate results in the I-unit firing. Furthermore, the ELL 203serves to reduce any timing jitter of phase information arriving fromthe T-unit, before then sending it to the TS 206 alongside the E-unitand I-unit spikes.

By comparing the positive zero crossing points within the beat signal103 (FIG. 1) to that of the reference signal 106 (FIG. 1), the TS 206 isable to determine whether the beat signal crossing points are leading orlagging the reference crossing points by outputting different spikingpatterns. The unit then performs a logical operation on the lead/lagphase information and the increasing/decreasing amplitude information,from which it decides whether to increase or decrease the dischargefrequency of the fish by stimulating the nucleus electrosensorius (nE+and nE−) 209 and ultimately the pacemaker nucleus (Pn) 212, which sets anew output frequency to be transmitted by the electric organ (EO) 215.The JAR decisions based on the multiple inputs from the P-unit and TS206 is summed up in the table of FIG. 2B, which compares the decisionwith its input amplitude and phase information, indicating an XNOR orXOR behavior.

FIGS. 3A and 3B illustrate the JAR logic operation, when the interferingfrequency f_(I) is lower than the reference frequency f_(E), and whenf_(I) is higher than f_(E), respectively. The beat signal (solid line303) and reference signal (dashed line 306) are shown with arrowspointing to different reference positive zero crossing points in whichthe beat signal is either leading or lagging the reference signal 306 inphase and the beat 303 is either increasing or decreasing in envelopeamplitude. FIGS. 3A and 3B show the TS operation principle given anoriginal reference frequency f_(E) of 100 MHz, with FIG. 3A portrayingthe scenario in which the interfering signal frequency f_(I) is at 80MHz, and FIG. 3B depicting the scenario in which the interferencefrequency f_(I) is at 120 MHz.

As can be seen in FIG. 3A, the nearest positive zero crossing point ofthe beat 303 is leading the nearest crossing point of the referencesinusoid 306 (f_(I) lead f_(E)) at the rising beat envelope and trailingat the falling beat envelop of the beat 303. The opposite trend (f_(I)lag f_(E)) is displayed in FIG. 3B for an interfering frequency f_(I)greater than the reference frequency f_(E). An optical implementation ofthe neural circuitry of FIG. 2A is described below, which makes use ofseveral different signal processing techniques towards determining beatamplitude information and interfering signal phase information.

Photonic Jamming Avoidance Response Circuitry

The basic understanding of the Eigenmannia JAR neural circuitry of FIG.2A serves as a rough guideline for developing a photonic equivalent.Considering that the reference frequency of most wireless transmittersis readily available from the local oscillator (LO), the referencesignal can be split and sent to the transmitter and to the T-unitdirectly. The ELL 203 no longer processes information from both theT-unit (phase coder) and P-unit (probability coder), thus the photoniccircuit can be divided into four major elements or subsystems forimplementation.

-   -   A P-unit, which can comprise an envelope detector and a circuit        that detects the rising edge of the beat envelope and emits an        optical pulse correspondingly.    -   A T-unit & ELL (or ELL/T-unit), which works to fire spikes at        every positive zero crossing point of the local oscillator's        output sinusoid.    -   A TS, which receives input directly from a receiver in the form        of two beating RF signals and from the T-unit & ELL in the form        of a train of spikes temporally indicating the positive zero        crossing points of the transmitter signal. The TS can use        optical effects to determine whether the interfering signal zero        crossing points are leading or lagging the reference signal's        zero crossing points and spikes accordingly.    -   A logic unit (gate), which interprets the phase and amplitude        information from the previous units and decides whether the        transmitter should increase or decrease its output frequency.        This information instructs the frequency adjustment unit to        control the signal generator and transmitter output is properly        adjusted.

FIG. 4 shows a simplified schematic diagram illustrating an example ofthe photonic JAR circuit comprising a receiver (Rx) 403 and transmitter(Tx) 406, the P-unit 409 for interpreting beat amplitude information,the ELL/T-unit 412 for indicating the positive zero crossing points of areference signal, the TS 415 for determining the phase of an interferingsignal relative to the reference signal, the logic unit 418 forinterpreting phase and amplitude information, and a local oscillator(LO) 421 for generating the reference signal. The photonic JAR circuitwas simulated using optical communications simulation software(OptiSystem), which demonstrated how the photonic JAR would operatearound 2.4 GHz in a wireless setting. A frequency adjustment (orcontrol) unit 427 can be used to provide feedback to the LO 421 foradjustment of the generated frequency of the reference signal. Thefollowing sections provide further detail for these subsystems.

P-unit—for the detection of rising edge of the beat signal envelope. TheJAR system of FIG. 4 can obtain a copy of its own RF signal fortransmission (reference signal) and an interfering signal from anunknown location by means of an antenna connected to the Rx 403. Beatingbetween the reference signal and the interference signal is firstobtained by the signal beating device 424 and sent to the P-unit 409where envelope detection is first performed to extract the envelope ofthe beat frequency. Then, the beat is converted to the optical domainwhere processing based on signal inversion, temporal offset and signaladdition can be performed. Ultimately, the P-unit 409 emits an opticalpulse at the rising edge of the envelope, and the resulting train ofpulses is sent to the logic unit 418 along a precisely determined lengthof optical fiber.

Referring to FIG. 5A, shown is a schematic diagram illustrating anexample of the photonic P-unit 409. The P-unit 409 can include twocontinuous wave (CW) optical sources 503, two electro-optic modulators(EOM), one positively biased 506 p and one negatively biased 506 n, anenvelope detector 509, a temporal delay 512, an optical amplifier 515, alength of highly nonlinear fiber (HNLF) 518 and a bandpass filter 521.In other implementations, different non-linear optical devices (e.g., asemiconductor optical amplifier (SOA), silicon wire waveguide orphotonic crystal fiber) can be utilized in place of the HNLF 518.

An envelope detector 509, in its most basic form, can be an RC circuitcomprising a diode, a resistor and a capacitor, in which the capacitorstores charge at the rising edge of the signal, while releasing itslowly on the falling edge through the resistor. The diode performsrectification, allowing current flow in just one direction, with the endresult being a signal that appears as a rectified sinusoid at the beatfrequency. FIG. 5B illustrates an example of envelope detection, wherethe beat of two signals at 2.4 and 2.6 GHz is shown by the dashed curve524, and the signal after envelope detection is indicated by the solidcurve 527. More advanced envelope detectors are available that operateover an array of different frequencies and are suited for multipleapplications. Due to the limitations of optical communicationssimulation software (OptiSystem) and the relatively simple nature of theenvelope detection 509, simulation of the envelope detection 509 was notneeded and the resulting signal was generated independently.

Referring back to FIG. 5A, the extracted envelope can be split into twobranches and converted to an optical signal through use of twoelectro-optic intensity modulators (EOM), one is biased at the positiveslope 506 p and one is biased at the negative slope 506 n so that anon-inverted copy and an inverted copy of the signal are obtained. TheEOM 506 is an electro-optic device that comprises two phase modulatorsto achieve amplitude modulation through interference. In an EOM 506, aninput optical signal is split between two branches, each containing aphase modulator. The phase modulator's index of refraction is a functionof the strength of an applied varying field, which in this case is theamplitude-varying envelope signal. Due to the varying index ofrefraction, the input continuous wave (CW) optical signal experiences atime-varying phase delay. Consequently, when the optical signals of bothbranches in the EOM 506 are recombined, they interfere and reproduce themodulated signal in the optical domain. Depending on the sign of anexternally applied bias voltage, the resulting modulated signal iseither upright or inverted.

The two branches of the P-unit 409 are slightly offset, such that aslight temporal delay 512 (e.g., of 1.5 ns) is experienced in the branchof the negatively-biased EOM 506 n before the two optical signals (e.g.,at 1551 and 1552 nm, respectively) are combined at a fiber coupler 530,where their intensities are added together. For example, two opticalsignals modulated by an envelope corresponding to a beating 2.4 and2.425 GHz signal are shown in FIG. 5C alongside the signal after thecoupler. FIG. 5D depicts the same information but for a 240 MHz beatsignal, corresponding to the beating between a 2.4 and 2.64 GHz signal.The optical signals from the positive-slope biased EOM 506 p (FIG. 5A)are shown as curves 533, the optical signals from the negative-slopebiased EOM 506 n (FIG. 5A) are shown as curves 536, and the signalsafter coupling by coupler 530 (FIG. 5A) is shown as curves 539.

The final element in the P-unit 409 of FIG. 5A is a 10 m length ofhighly nonlinear fiber (HNLF) 518, in which the nonlinear optical effectof four-wave mixing (FWM) is utilized to produce a pulse at the risingedge of the beat. As the signals (curves 539 of FIGS. 5C and 5D) aftercoupling by coupler 530 do not maintain the same pulse shape over therange of beat frequencies, further processing can be used to emit aproper pulse. The processing technique used, FWM, occurs when at leasttwo signals at different optical frequencies, v₁ and v₂, propagatethrough nonlinear media and produce two additional frequency componentsat v₃=v₁−(v₂−v₁) and v₄=v₂+(v₂−v₁), where v₂>v₁, assuming that theinputs are phase-matched. Because of the high optical power needed forFWM, the effect only occurs at the rising edge of the beat, where thetwo optical inputs are of sufficiently high power. The optical amplifier515 preceding the HNLF 518 is initialized for lower frequency beats,which correspond to a small separation between the reference andinterfering signal, such that output of the P-unit 409 is a spiketemporally aligned to the crossing point between the two opticalsignals. Near this crossing point, the power of each signal issufficiently high to induce a noticeable FWM effect, and a 0.3 nmbandpass filter 521 (e.g., centered at 1550 nm) passes the FWM product.FIG. 5E illustrates an example for a 25 MHz beat envelope.

For higher beat frequencies, a fixed temporal delay 512 (FIG. 5A) of,e.g., 1.5 ns shifts the inverted signal further along the rising edge ofthe envelope in terms of relative phase. This leads to a minor relativephase shift in the output of the P-unit 409 (FIG. 5A) for higherfrequency signals as the power needed for FWM is effectively satisfiedat all times corresponding to the peak amplitude of the inverted signal.This trend is further supported by the fixed bias current supplied tothe amplifier 515. FIGS. 5E-51 show the output of the P-unit 409 forbeat frequencies of 25, 50, 100, 200 and 240 MHz, respectively.

In FIGS. 5E-51, the beat modulated optical signals 533 and 536, and theFWM products 542, are shown for interfering and reference signalseparations of 25, 50, 100, 200 and 240 MHz, respectively. In each ofthe figures, the vertical axis corresponds to the P-unit spikeamplitude, and the initial envelopes 533 and 536 are scaled down forreference. In actuality, the average power of the coupled inputs 533 and536 was approximately 7 dBm for all tested beat frequencies, and theaverage output powers of the P-unit 409 were −12, −10.3, −7.8, −7.7, and−7.5 dBm for the 25, 50, 100, 200, and 240 MHz beats, respectively.

Considering that operation of the photonic JAR circuitry is not neededfor frequencies that are sufficiently far apart to avoid interference, a250 MHz bandpass filter can be positioned after envelope detection,preventing the modulation of any beat signal larger than this frequencyonto the optical carriers. In such a scenario the two optical signalsthen fail to initialize FWM, and the P-unit 409 entirely stopsspiking—disabling the JAR operation. FIG. 5J illustrates the effect of a280 MHz envelope before and after a low pass filter, and FIG. 5K showsthe output of the P-unit.

The 250 MHz low-pass filter prevents the P-unit 409 from spiking at therising edges of a 280 MHz beat. In FIG. 5J, the 280 MHz beat isindicated by curve 545 prior to filtering and the straight line 548represents the signal after filtering. As shown in FIG. 5K, the P-unit409 does not spike as FWM fails to occur. Evidently, conversion of theenvelope to the optical domain fails, resulting in no output from theP-unit 409. Also, the bandpass filter prevents any issues related to thefixed delay line from arising. For instance, for beats larger than about333 MHz the delay would align the peak powers of the two opticalsignals, resulting in spikes corresponding to the beat's peak. Suchbehavior would result in ambiguity between the beat's rising and fallingedge, and the P-unit 409 would output a spike at the beat's falling edgefor frequencies just larger than about 333 MHz.

Ultimately, FIGS. 5E-51 illustrate the P-unit's ability to properlyspike at the rising edge of the beat signal for frequency separationsranging from 25 to 240 MHz, based on a fixed temporal delay, FWM, andoptical filtering of one of the FWM generated frequencies. The output'stendency to shift towards the beat peak at higher frequencies, such as240 MHz, results in a minor temporal misalignment between inputs at thelogic unit 418 of the JAR circuitry of FIG. 4.

T-unit & ELL—for the detection of the reference signal's positive zerocrossing points. Referring to FIG. 6A, shown is a schematic diagramillustrating an example of a T-unit & ELL (or ELL/T-unit) 412 of FIG. 4,with a CW source 603 modulated by an EOM 606 receiving the referencesignal directly from the LO 421 (FIG. 4). A nonlinear optical loopmirror (NOLM) based thresholder 609 performs amplitude clamping, asillustrated by the inset 612 of FIG. 6A. An optical amplifier 615receives the output of the thresholder 609 and supplies a length ofnonlinear fiber (HNLF) 618 connected to a bandpass filter 621. In otherimplementations, different non-linear optical devices (e.g., a SOA,silicon wire waveguide or photonic crystal fiber can be utilized inplace of the HNLF 618.

The ELL/T-unit 412 receives a copy of the reference signal directly fromthe LO 421 and, because of the fixed distance between the transmittingand receiving antennae, the length of the fiber from the LO 421 can beset so that the reference signal is in phase with the reference signaltransmitted and received by the antennae at a later point in the overallsystem. The system can then locate the positive zero crossing points ofthe reference signal by means of amplitude clamping, self-phasemodulation (SPM), and spectral filtering.

For the first step in this process, the reference sinusoid passesthrough an optical thresholder 609, similar to those implemented byphotonic leaky integrate-and-fire (LIF) neuron models, with transferfunctions similar to that of FIG. 6B. FIG. 6B shows the DREAMthresholder transfer function, comparing input and output powers of 10ps pulses. A similar device would be used for amplitude clamping. Theoptical thresholder 609 works to suppress signals below a set powerthreshold and pass those beyond the threshold with a sigmoidal transferfunction. Consequently, the sinusoid is clamped in such a way that itscrests and troughs are flattened, while the rising and trailing edgesare made steeper. To surpass simulating the amplitude clamping effect,non-return-to-zero (NRZ) signal of appropriate width are generatedelectronically at the same rate as the reference signal frequency andconverted to the optical domain using another EOM to modulate the signalonto a 1550 nm continuous wave.

The leading edges of these NRZ signal can be roughly aligned in timewith the positive zero crossing points of the reference sinusoid, and afiltering technique similar to NRZ-to-PRZ (pseudo-return-to-zero) signalconversion can be used to extract these edges. Considering the nonlinearproperties of fiber, an NRZ signal can induce a minor red shift in itsleading edge and a minor blue shift in its trailing edge, and NRZ-to-PRZconversion is therefore achieved by filtering out the central wavelengthof the signal, leaving narrow pulses at the rising and trailing edges ofthe original NRZ signal. By performing a similar technique, but insteadjust filtering out the longer wavelength end of the NRZ signal'sspectrum, extraction of exclusively the leading edge is possible.

The manifestation of this process within the photonic JAR circuitinvolves first sending the optical NRZ signal through another length ofHNLF 618, in which SPM occurs, broadening the spectrum andsimultaneously inducing further red shifts and blue shifts. SPM is anonlinear phenomenon describing how a medium's index of refractivechanges proportionally to input optical intensity, resulting in a shifttowards the longer wavelengths at the leading edge and a shift towardsshorter wavelengths at the trailing edge of a pulse. Then, the signalpasses through an optical bandpass filter 621 (e.g., with a 0.3 nmbandwidth centered at 1550.52 nm) at the longer wavelength side of theoriginal pulses, and output of the ELL/T-unit 412 is acquired. Whileextraction of the zero crossing point can be done directly with thereference sinusoid, the use of thresholder to convert the sinusoid intoan NRZ signal does enhance the SPM effect for generating a stronger andsharper zero-crossing pulses.

FIGS. 6C and 6D show the frequency and time domain simulation outcomefor this technique performed on a 2.4 GHz NRZ signal. FIG. 6C shows theNRZ signal spectrum 624 and the filtered leading edge spectrum 627. InFIG. 6D, the 2.4 GHz positive zero crossing point pulses shown in curve630 with a 2.4 GHz reference sinusoid shown as curve 633. A train ofnarrow optical pulses, in synch with the positive zero crossings of thereference signal and an average power of −7.8 dBm, is emitted and sentto the TS unit 415 (FIG. 4), where phase information is processed.

TS unit—for determine the phase of the beat signal relative to thereference signal. Referring to FIG. 7A, shown is a schematic diagramillustrating an example of a TS unit 415 of FIG. 4. FIG. 7A shows thebeat signal 709 modulated onto an optical carrier (at λ₀) from CWoptical source 703 by an EOM 706 and the pulse train 712 (at λ₁)generated from the ELL/T-unit 412. Both inputs 709 and 712 are coupledtogether by a fiber coupler 715 and sent to a semiconductor opticalamplifier (SOA) 718 and a bandpass filter 721 (at λ₁) passes the pulsetrain. Ultimately, the output of the TS unit 415 indicates at whichpoints the interfering signal is lagging behind that of the reference.

The TS unit 415 in FIG. 7A receives an input from the ELL/T-unit 412 anddirectly from the receiver 403 via the antenna and the signal beatingdevice 424. The TS 415 includes a SOA 718, which is utilized in a mannerrepresentative of an integrator in a photonic neuron, and serves toindicate at which times the interfering signal lags the reference signalin phase. A electro-absorption modulator (EAM) based photonic neuronintegrator can also be used. First, the beating signals are converted tothe optical domain by an EOM 706, modulating the interference patternonto a 1555 nm optical carrier. Then, this new optical signal with anaverage power of −8.7 dBm is sent to the SOA 718 alongside the positivezero crossing pulses (attenuated to an average power below −25 dBm) at1550 nm, which act similarly to the sampling pulses of an LIF photonicneuron. A 2 nm bandpass filter 721 centered at 1550 nm can be used topass the sampling pulse train 712, whose pulses' amplitudes are alteredby cross-gain modulation (XGM).

FIGS. 7B-7E shows both inputs for two different scenarios, in which theinterfering signal is either 100 MHz above (FIG. 7B) or below (FIG. 7C)a 2.4 GHz reference signal. In the comparison of the TS inputs andoutputs for 100 MHz separations between reference and interferingfrequencies, the zero crossing pulses are show as 724 and beat signal as727. FIG. 7B shows the TS inputs for a 2.4 GHz reference signal and a2.5 GHz interfering signal and FIG. 7C shows the TS inputs for a 2.4 GHzreference signal and a 2.3 GHz interfering signal. The TS outputs arepictured in the two scenarios, showing that the crossing point pulsesare amplified the most when the beat signal is lagging behind thereference signal. FIG. 7D shows when the beat frequency is greater thanthe reference frequency by 100 MHz, and FIG. 7E shows when theinterfering frequency is less than the reference frequency by 100 MHz.Time intervals at which the beat signal is lagging and leading areindicated by the red text and blue text, respectively.

As can be seen from FIG. 7B, the zero-crossing pulses are temporallycloser to the individual peaks in the beat signal at the fallingenvelope of the beat than they are at the rising envelope. The exactopposite relationship is observable in FIG. 7C, which corresponds to thescenario in which the interfering signal is lower in frequency than thereference signal. When filtering out the zero-crossing pulses by meansof an optical bandpass filter 721 (FIG. 7A) with a 2 nm bandwidthcentered at 1550 nm, different outputs are observed for the twoscenarios as a result of the SOA's gain dynamics.

In FIG. 7B, the pulses and peaks at the beat's rising envelop are spacedout such that the SOA 718 (FIG. 7A) has adequate time to recover betweeninputs, such that the beat signal will not deplete the gain experiencedby the pulses, and the output pulses are thus amplified to a furtherextent at these times; however, at the falling envelope of the beat, thepulses are temporally closer to the small peaks and thus the gainexperienced by the pulses are depleted by the beat signal peaks, thusexperience minimal amplification. As such, the output depicted in FIG.7D results. When the interfering signal is lower in frequency, however,the output shown in FIG. 7E occurs due to the now increased spacingbetween pulses and beat peaks at the beat's falling edge and decreasedspacing at the rising edge. The spiking behavior directly relays phaseinformation of the beat signal. For instance, when the interferingfrequency is lower than the reference frequency, the rising edge of theresulting beat envelope always occurs when the beat signal's positivezero crossings are leading the reference's crossing points, and thefalling edge always corresponds to the beat signal's crossings laggingbehind that of the reference's. The opposite behavior is observed whenthe interfering frequency is higher than the reference frequency.Consequently, this TS implementation 415, which is of a basic design, isreminiscent of the very first LIF photonic neuron and outputs a “1” whenthe beat signal is lagging behind the reference signal and an effective“0” when the beat signal leads the reference signal.

The TS unit 415, while successful in conveying the necessary phaseinformation, does not produce the most ideal output. For instance, someof the techniques used by more complete photonic neuron models could beutilized to clean up the TS output. Thresholding by means of a nonlinearoptical loop mirror (NOLM) can suppress the zero level of the signal,which can make for easier signal processing at later stages of thephotonic JAR circuit, and an inverter and second thresholder couldfurther serve to increase the disparity between ones and zeros whilereversing the logic of the output. The device would then output spikeswhen the interfering signal leads the reference signal and transmitnothing when the interfering signal lags behind the reference signal.

Logic unit—for determination of output frequency adjustment. The logicunit 418 of the photonic JAR circuit processes the amplitude and phaseinformation acquired from the P-unit 409 and the TS 415, making a finaldecision as to whether or not the transmission frequency is in need ofadjustment and in what direction. Operating as a basic XOR or XNOR gate,the decision from the logic unit 418 is summarized in the table of FIG.8, which lists the decisions based on amplitude and phase information.For this system, a spike from the P-unit 409 marks the rising edge ofthe beat envelop, and a strong spike from the TS 415 indicates times atwhich the beat signal is lagging behind the reference signal; however,the logic of the TS 415 can easily be reversed with additional photonicneural components. The inputs of the logic unit 418 are both in theoptical domain, and several approaches for XOR or XNOR logic can beapplied with this system. With beat frequencies in the MHz range,electrical processing can be used instead of optical processing.

FIGS. 9A-9H show the inputs for several different interference scenariosafter conversion to the electrical domain by photodetectors and passagethrough 500 MHz low-pass filters. The logic inputs are shown for a 2.4GHz reference and an interfering signal of 25 MHz higher in FIG. 9A, of25 MHz lower in FIG. 9B, for 100 MHz higher in FIG. 9C, for 100 MHzlower in FIG. 9D, for 200 MHz higher in FIG. 9E, for 200 MHz lower inFIG. 9F, for 240 MHz higher in FIG. 9G, and for 240 MHz lower for FIG.9H. The filtering was implemented because of the different nature ofP-unit 409 and TS 415 outputs 903 and 906, respectively, with the P-unit409 transmitting nanosecond-width pulses at MHz repetition rates, andthe TS 415 emitting ultrashort, picosecond width pulses at GHzrepetition rates.

Without adjusting the system parameters for any of the simulationtrials, the P-unit 409 and TS 415 exhibit the proper spiking behaviorfor the situations in which the interfering signal was up to 240 MHzhigher or lower in frequency than a reference signal at 2.4 GHz. It wasnoted that, in some trials, the amplitudes of the TS 415 spikes 906 andP-unit 409 spikes 903 varied by as much as 300 μW. While the“zero”-level of the TS output 906 was maintained at around 100 μW forall trials, there exists variance in pulse widths between the logicinputs. Particularly at the lower frequencies, the logic inputs did notappear to be temporally aligned, and the pulse shape indicates that weakFWM was occurring along the falling edge of the beat as well.

More precise adjustment of the system parameters can eliminate ormitigate several of these issues. Adding a fixed attenuator to theP-unit 409 or fine tuning of the TS inputs may result in a more evenmatch in amplitude for the logic inputs. Attenuation and amplificationof the crossing point pulses and beat signal TS inputs, along withoptimization of the SOA driving current or thresholding and inversiontechniques, can significantly alter the TS output. Also, the P-unitoutput at low frequencies may be minimized by optimizing the drivingcurrent to the system's amplifier preceding the HNLF. By determining theideal bias current, the minor leading spike, as seen in FIG. 5Ecorresponding to the crossing point of the negative envelope with thepositive envelope's falling edge, could be minimized if the two HNLFinputs are not of sufficient power to induce FWM. Lastly, the lack oftemporal alignment for the lower frequencies is simply a product ofsimulation, as the software employed has a difficult time accuratelyrepresenting a wide range of frequencies in a single program.

Also, the results describe how a system with a reference signal fixed at2.4 GHz would respond to different interfering frequencies, when inactuality the reference frequency can be adjusting depending on thelogic circuit output. The decision to keep the reference frequency fixedand observe the circuit's response to difference interfering frequencieswas made due to simulation limitations. The OptiSystem simulationoperates on a predefined bit rate upon which the generation of the NRZpulses, and thus the positive zero crossing point detection, depends on.If the reference frequency were adjusted from 2.4 GHz by severalmegahertz, the fixed bit rate of the simulation would result in anerroneous depiction of the reference signal's crossing points. Despitethis shortcoming, the simulations indicate the ability of the photonicJAR system to properly respond to a range of frequency differences, andadjustment of the interfering frequency directly parallels theadjustment of the reference frequency.

The photonic JAR circuit of this disclosure produces the necessaryspiking patterns on which a basic XOR logic gate can operate. For thisbasic model, a fixed low-pass filter at 250 MHz prevents the P-unit 409(FIG. 4) from spiking for frequencies sufficiently far from thereference frequency, providing a potential stopping condition for theJAR circuit. The disclosed system does not make use of any stronglyfrequency-dependent components, allowing for operation over a widefrequency range, with previous simulations successfully operating atfrequencies around 10 GHz, and both electrical and optical processingtechniques can be implemented in the final decision system. Also, allprocessing techniques used do not depend on the phase of the interferingsignal, ensuring that the location of the interfering signal'stransmitter is inconsequential to the circuit's outcome. Otherimplementations can include complete, autonomous operation of the JARcircuit, including simulation of NOLM thresholding towards generatingthe positive zero crossing pulses, envelope detection for rising edgedetection of the beat, a XOR/XNOR logic operation on the P-unit and TSoutputs, and/or an adaptive feedback response that properly adjustsand/or appropriately stops adjusting the reference frequency based onthe logic unit's decision. In addition, the disclosed system can beutilized for signals of various modulation formats beyond sinusoids atfixed amplitudes, including a variety of inputs present in a real worldenvironment.

Experimental Results of Photonic JAR

The Eigenmannia performs each task through spike processing, thereforemultiple neurons and units may be needed for encoding certain functions.From an engineering point of view, it may be redundant to implement twoseparate devices for the same function. Thus, in the opticalimplementation of JAR, the circuitry was simplified and make use ofphotonic phenomena for an efficient photonic JAR. In a radar system, thetransmitter (Tx) carrier signal corresponds to the Eigenmannia referencesignal. As shown in FIG. 4, the Tx signal can be obtained from the localoscillator (LO) 421, while the interfering signal can be obtained fromthe receiver (Rx) 403, e.g., after a self-interference cancellationsystem. During the experiments, a Tx signal frequency in a range fromhundreds of MHz to 10 GHz was used, while the interfering signal variedfrom 240 MHz above the TX signal frequency to 240 MHz below the Txsignal frequency. Both the simulation results and experimental resultsillustrated the JAR behavior. The experimental version of the JAR isdesigned for performance optimization and potential device integration.

The first unit in the JAR system is a P-unit 409, which is used for thediscrimination of increasing or decreasing amplitude in the beatenvelope. Referring to FIG. 10A, shown is a schematic diagramillustrating an embodiment of the photonic P-unit 409, which is based ontemporal offset and subtraction of signals. The optical P-unit 409 wasimplemented by performing electrical-to-optical conversion of the beatsignal envelope 527 through the use of an electro-optic modulator (EOM)506, as illustrated in 1003. Due to the close frequency differencebetween the reference signal and interfering signal (<hundreds of MHz),envelope detection of the beat signal 524 can be done electrically.

The optical envelope signal 527 was combined with a continuous wavelaser output 1006 at a different wavelength, and launched into asemiconductor optical amplifier (SOA) 519 for cross gain modulation,such that both a non-inverted copy 1009 and an inverted copy 1012 of thebeat signal envelope 527 were obtained. Optical bandpass filters 522were used to separate the two signals, with the inverted signal 1012temporally delayed 513 by a fixed amount of time (e.g., 1.5 ns) andcombined with the non-inverted signal 1009 by an optical coupler. Due tothe temporal offset, the combined signal 1015 is at peak power duringthe increasing envelope of the beat signal envelope 527. Therefore, arising envelope can be identified by a “bit 1” at the P-unit output. Inthe experiment, the frequency sensitivity (the maximum frequencydifference between the Eigenmannia reference signal and the interferingsignal for enabling JAR) was set to be 150 MHz. The results shown herecorrespond to a reference signal at 1 GHz while the interfering signalwas at 1.01 GHz. The input and output waveforms of the P-unit 409 areshown in FIG. 10B, with the lower curve as the beat signal envelope 527and the upper curve as the P-unit output 1015. It can be seen that theP-unit 409 returned a level “1” during the increasing amplitude portionof the beat signal (shaded areas), but not during the decreasingportion.

The next unit is the ELL/T-unit 412, which is responsible foridentifying the positive zero crossing points in the Tx signal. Anembodiment for positive zero crossing detection is illustrated in FIG.10C. The ELL/T-unit 412 received a copy of the Tx signal directly fromthe LO 421, which was converted to the optical domain reference signalthrough electrical to optical conversion. The optical reference signalwas then passed through a SOA 619 (e.g., an optical nonlinear medium toexperience self-phase modulation) such that the rising edge of thereference signal induced a shift to the longer wavelength direction andthe falling edge induces a shift to the shorter wavelength direction(spectral broadening). Then, the positive zero crossing region of thesinusoidal signal was extracted based on offset filtering 622 of thespectrally shifted output. FIG. 10D shows the output pulses 630 from theELL/T-unit 412, which are aligned with the positive zero crossing pointsof the sinusoidal Tx signal 633, indicating that the positive zerocrossing points were successfully identified by the ELL/T-unit 412.

Next, the TS unit 415 receives the positive zero crossing informationfrom the ELL/T-unit 412 along with a copy of the beating signal envelopebetween Tx (reference) and interfering signals, and determines whetherthe beat signal is leading or lagging the Tx signal. FIGS. 10E and 10Fshow examples of the inputs and outputs of the TS unit 415 for 10 MHzseparations between a 1 GHz Tx signal and the interfering signal. Forexample, if the positive zero crossing points are aligned with thepositive amplitude of the beat signal, that means the beat signal isleading the Tx signal; however if the positive zero crossing pulses arealigned with the negative amplitude of the beat signal, the beat signalis lagging the Tx signal. To realize this, the optical version of thebeat signal 524 (FIG. 10A) is combined with the zero crossing pulses 630(FIG. 10D) and launched into a semiconductor optical amplifier (SOA) forcross-gain modulation. If the positive zero crossing pulses are slightlybehind but very close temporally to the beating signal peaks, the outputpulses after the SOA will be weakened; however, if the positive zerocrossing pulses are in front of the beat signal peaks, the output pulseswill be strengthened.

To exemplify the process experimentally, two different scenarios inwhich the interfering signal was either 10 MHz below or above a 1 GHzreference signal (f_(I)<f_(E) or f_(I)>f_(E)) are shown in FIGS. 10E and10F, respectively. The plots indicate the beat signal 524 and the output1018 of the TS unit 415. As can be seen from FIG. 10E, the zero crossingpulses were suppressed during the first half (shaded region) of the beatsignal 524, while the pulses were strengthened during the second half ofthe beat signal 524. This condition corresponds to the interferingsignal being lower in frequency than the Tx (reference) signal. Theexact opposite relationship is observable in FIG. 10F, which correspondsto the interfering signal being higher in frequency than the Tx signal.Here, the TS unit output pulses 1018 with a stronger amplitude duringthe first half (shaded region) of the beat signal 524 mean that thephase of the beat signal is lagging the Tx signal, while pulses withweaker amplitude during the second half correspond to leading phase. Thepulse amplitude directly provides phase information of the interferingsignal.

The final units of the JAR system is the logic unit 418 and frequencyadjustment (or control) unit 427 (FIG. 4), which processes the amplitudeand phase information acquired from the P-unit 409 and the TS unit 415,making a final decision as to whether or not the Tx frequency is in needof adjustment and in what direction. The control unit 427 will regulatethe control to the LO signal generator 421 by taking into account theJAR logic decision. For this system, a peak from the P-unit 409 marksthe rising edge of the beat signal envelope 527, and strong pulses 1018from the TS-unit 415 indicate times at which the beat signal is laggingbehind the Tx (reference) signal; thus an XOR logic operation can beutilized by the logic unit 418 to return a “1” for an increase infrequency and a “0” for a decrease in frequency. The XOR operation canbe implemented either optically or electrically, depending on the actualsystem requirements. In this study, the JAR process was enabled onlywhen the interfering signal was within the hundreds of MHz range of thereference signal. Consequently, the resultant logic operation wascarried out electrically due to the low frequency of the signal. Alow-speed photodetector with a hundreds of MHz bandwidth was used forconverting the optical signal to the electrical domain as well asband-limiting the signals. FIG. 10G illustrates the logic unit 418 andcontrol unit 427 used for testing. The circuitry included a XOR gatethat received phase information from the TS unit 415 and amplitudeinformation from the P-unit 409 to determine the frequency shiftingdirection, a bias-to provide a DC offset to the integrator for properoperation, while an integrator was used to hold the driving voltage ofthe LO 421. The XOR gate can be enabled to provide an output when thejamming signal frequency is within a defined range of the Tx (reference)frequency (e.g., equal to and/or less than a predefined frequencydifference), and no output when the jamming signal frequency is outsidethe defined range. FIGS. 10H and 10I show the corresponding inputs tothe XOR logic, in different scenarios. The amplitude information 1015corresponds to the peaks from the P-unit 409 marking the rising edge ofthe beat signal envelope 527, while the envelope 1021 of the phaseinformation 1018 is detected using an envelope detector, thatcorresponds to the detected strong pulses from the TS unit 415, whichindicate times at which the beat signal is lagging behind the Tx signalor leading the Tx signal.

FIG. 10H shows the case where the interference signal is at a lowerfrequency than the Tx signal (f_(I)<f_(E)), with a frequency differenceof 10 MHz. While FIG. 10I shows the case where the interference signalsare higher in frequency than that of the Tx signal (f_(I)>f_(E)), with afrequency difference of 10 MHz. According to the XOR logic operation,FIG. 10H produces a high logic level at the output of the logic unit418, while FIG. 10I produces a low level at the output. A high levelindicates that an increase in frequency is needed, while a low levelindicates that a decrease in frequency is needed. The logic output isthen provided to the control unit 427 to control the Eigenmannia (Txsignal) signal generator (LO 421) accordingly. The above experimentdemonstrates that the optical implementation of the JAR circuit iscapable of operating at a much higher frequency and wider frequencyrange than that of its biological counterpart.

The decision made by the logic unit 418 and control unit 427 will belaunched back to the LO 421 of the Tx, such that a correspondingfrequency change can be made automatically to avoid jamming. In thisexperiment, the JAR circuit was triggered when the jamming signal was180 MHz or less away from the Tx (reference) frequency. This can beadjusted by controlling the bandwidth of the low pass filter before thefirst envelope detector. FIGS. 10J-10L show the photonic JAR system inaction. FIG. 10J illustrate when the jamming signal is >150 MHz from theTx frequency, so that the JAR system will not be enabled. When thejamming signal approaches the Tx frequency (<180 MHz away), the JARsystem can be enabled. FIG. 10K illustrates that the Tx frequency ispushed to a higher frequency. As the jamming signal keeps moving to thehigher frequency, the Tx frequency will be pushed to an even higherfrequency by the JAR system. In this way, the interfering signal cannever get closer than 180 MHz from the Tx frequency, as illustrated inFIG. 10L.

Photonic Jamming Avoidance Response System Design

Referring next to FIGS. 11A and 11B, which show schematic diagramsillustrating another example of a photonic JAR system design. As shownin FIG. 11A, the JAR system comprises four units referred to as azero-crossing point detection (ZeroX) unit 1103, a phase detection unit1106, an amplitude unit 1109 and a logic unit 1112. In the photonic JARdesign of FIG. 11A, a reference signal 1115 is provided by a localsource (e.g. for transmission) and the jamming (or interfering) signal1118 is received from an external source.

The reference signal 1115 is directed to the Zero-crossing pointdetection unit 1103 for the identification of positive zero-crossingpoints in the reference signal 1115, which will later be used as a phasereference for the beat signal. The ZeroX unit 1103 generates a spike ateach positive zero-crossing point of the reference signal 1115, asillustrated by the pulses in inset (i) of FIG. 11A. The reference signal1115, S_(R)=sin(2πf_(R)t), and the jamming (or interference) signal1118, S_(J)=sin(2πf_(J)t), are combined using an RF combiner 1121 wherea beat signal, S_(B)=sin(2πf_(R)t)+sin(2πf_(J)t), results as shown ininset (iii) of FIG. 11A.

The beat signal and the zero-crossing spikes are directed to the phasedetection unit 1106 to determine when the beat signal is phase leadingor lagging the reference signal 1115. The phase information canrepresented by the modulated amplitude of the zero-crossing spikes atthe output of the phase detection unit 1106 as shown in inset (ii) ofFIG. 11A. Next, amplitude rising/falling information of the beat signalis extracted by directing the envelope of the beat signal to theamplitude unit 1109, as shown in insets (iv) and (v) of FIG. 11A.Finally, the phase and amplitude information are fed to the logic unit1112 in which the JAR system can determine how, if necessary, to controlthe reference signal 1115.

Zero-Crossing Point Detection Unit. The photonic implementation of JARcan be based on the use of semiconductor optical amplifiers (SOA),starting with the zero-crossing point detection unit (ZeroX unit) 1103.The goal of the ZeroX unit 1103 is to identify the positivezero-crossing points of the reference signal 1115, which will later beused as the phase reference in the phase detection unit 1106. As shownin FIG. 11B, the sinusoidal reference signal 1115 from a voltagecontrolled oscillator (VCO) is launched to a clock divider (CD)operating in a divide-by-1 configuration before being amplitudemodulated (e.g., using an electro-optic intensity modulator (EOM 1))onto an optical carrier (e.g., at 1549.35 nm) from a distributedfeedback laser (DFB 1).

After launching to the clock divider 1127, the reference signal 1115becomes square-like, which enhances the detection later at the SOA. Theoptical reference signal 1115 is amplified by an erbium doped fiberamplifier and is launched to the ZeroX unit 1103, which comprises an SOA(SOA 1) and an optical bandpass filter (OBPF 1). Self-phase modulationoccurs in the SOA 1 at each rising and falling edge of the square-likereference signal, and broadening (e.g., about 0.4 nm) at the longerwavelength side (the rising edge side) can be observed. The OBF 1 (e.g.,at 1549.50 nm) can be used to select the desired red shifted portion,where the resultant peaks align exactly at the positive zero crossingpoints of the reference signal 1115.

FIG. 11C shows the sinusoidal reference signal at various frequenciesf_(R) 1203 and the resultant positive zero crossing pulses 1206generated from the ZeroX unit 1103. The ZeroX unit 1103 works well forreference signals from hundreds of MHz to 20 GHz due to the fast 25-psrecovery time of the SOA.

Phase Detection Unit. The goal of the phase detection unit (phase unit)1106 is to determine whether the instantaneous phase of the beat signalis leading or lagging the phase of the reference signal. As shown inFIG. 11B, the phase unit 1106 takes as inputs the zero-crossingreference, pulsed output 1206 from the ZeroX unit 1103 and the beatsignal between the reference signal 1115 and jamming signal 1118. Thebeat signal is amplitude modulated (EOM 2) onto an optical carrier froma distributed feedback laser (DFB2) (e.g., at 1552.57 nm). The phaseunit 1106 has the same structure as the ZeroX unit 1103, which comprisesan SOA (SOA 2) and an optical bandpass filter (OBPF 2), but a differentoptical phenomenon, cross-gain modulation (XGM), is utilized.

The OBPF 2 can be used to extract the zero-crossing pulses at the outputof the SOA 2 after experiencing cross-gain modulation. The beat signalacts as the pump to introduce a gain change in the SOA 2 through gaindepletion, and the zero-crossing reference pulses act as the probe toexperience the gain change, if there is any. Since the beat signal isoscillating like a sinusoidal wave with modulated amplitude, eachoscillating cycle comprises positive periods and negative periods. Theoptical power of the beat signal can be adjusted such that only thepositive period is strong enough to introduce significant gaindepletion, while the “negative” period will be too weak for gaindepletion.

Since the SOA 2 has an exponential gain recovery curve, the strength ofcross-gain modulation strongly depends on the temporal spacing andsequence between the pump (beat) and probe (zero-crossing pulse)signals. If the instantaneous phase of the beat signal is leading thereference signal phase, the positive zero-crossing point of thereference signal will pass through the SOA 2 after the arrival of thepositive period. Thus, zero-crossing reference pulses experiencesignificant cross-gain modulation, diminishing their amplitudes. On theother hand, if the instantaneous phase of the beat signal is lagging thereference signal phase, the positive zero-crossing point of thereference signal will align with the “negative” period of the beatsignal and pass through the SOA 2 before the arrival of the positiveperiod. Thus, the zero-crossing reference pulses do not experience thesame level of SOA gain depletion and increase in amplitude. The SOA 2 isdriven such that the recovery time is on the order of half of thereference signal period but much shorter than the beat signal frequency|f_(R)−f_(J)|. Thus, independent cross-phase modulation occurs duringeach beat signal oscillation but not across multiple oscillation cycles.

FIG. 11D shows the waveforms of the input beat signal 1209 and theoutput zero-crossing reference pulses 1212 after cross-gain modulationin the SOA 2. An envelope detector (ED) can be used to extract theenvelope of the modulated zero-crossing reference pulses 1215. Thereference signal can be at 1 GHz, while the jamming signals can be 10MHz, 50 MHz and 100 MHz above the reference signal frequency(f_(R)>f_(J) as illustrated in plots i, ii and iii, respectively) orbelow the reference signal frequency (f_(R)<f_(J) as illustrated inplots i, ii and iii, respectively). Pulses with amplitudes above thedashed line can be regarded as high amplitude (considered as a binary“1”)—representing that the beat signal is phase lagging, while pulseswith amplitude below the dashed line can be regarded as low amplitude(considered as a “0”)—representing that the beat signal is phaseleading.

FIG. 11E illustrates the relationship between the reference f_(R)(dashed) and the beat signal (solid), when (a) f_(R) is at a higherfrequency than f_(J), and (b) f_(R) is at a lower frequency than f_(J).The solid curve following the peaks of the beat signal represents onecycle of the beat signal envelope. The positive zero-crossing points ofthe beat signal (crosses) travel around the positive zero-crossingpoints of the reference signal (circles) along the beat envelope period,implying that the phase of each oscillation in the beat signal changesover the envelope period. In the top plot of FIG. 11E, wheref_(R)>f_(J), the phase of the beat signal is lagging that of thereference signal at the falling edge of the beat signal envelope; whileit is leading the phase of the reference signal at the rising edge ofthe envelope. On the other hand, in the bottom plot of FIG. 11E, wheref_(R)<f_(J), the opposite relationship is observed—at the falling edgeof the envelope, the phase of the beat signal is leading that of thereference signal; while it is lagging the phase of the reference signalat the rising edge of the envelope.

The instantaneous phase of the beat signal changes between leading tolagging the reference signal phase, and a variation in the resultingreference pulse amplitudes are consequently expected. Therefore, whenexamining FIG. 11D where the f_(J)>f_(R), a leading in phase response(“1”) is always observed at the beat envelope's falling edge, while alagging in phase response (“0”) is always observed at the envelope'srising edge. When f_(J)<f_(R), an opposite relationship is observed—alagging in phase response (“0”) is always observed at the falling edge,while a leading in phase response (“1”) is always observed at the risingedge. The above observation match precisely with the predictions,illustrating the accurate identification of the phase relation betweenthe beat and reference signals.

Amplitude Unit. Although the phase and amplitude relationship areobservable with an oscilloscope, a unit capable of identifying therising and falling edge of the beat envelop is needed to ensure anautonomous JAR response, which is an important advantage of JAR overother manual anti-jamming schemes. Thus, the goal of the amplitude unit1109 is to return a different value for the rising and falling edge inamplitude portions of the beat envelope, e.g., a positive value outputto indicate a rising amplitude and a negative value output to indicate afalling amplitude. The principle of the amplitude unit 1109 can be basedon temporal offset and signal subtraction, illustrated in inset (iv) ofFIG. 11A. Signal subtraction can be considered as signal addition withits complement, i.e. A−B=A+(−B). Therefore, when a small temporal delayis introduced in the negative version of the signal, the resultingsignal will not be zero. Instead, a rising amplitude will result in apositive value while a falling amplitude will result in a negativevalue.

To implement rising/falling amplitude detection in the amplitude unit1109 of FIG. 11B, the beat signal envelope can first be extracted by anenvelope detector (ED), which can be accomplished using RF electronicsdue to the beat's low frequency |f_(R)−f_(J)|. The beat signal envelopecan be amplitude modulated (EOM 3) onto an optical carrier from adistributed feedback laser (DFB 3) (e.g., at 1553.33 nm). Since anoptical signal cannot be negative, an inverted copy of the beat signalenvelope can be used instead—which will be the same except with apositive offset value. Cross-gain modulation between a continuous wavelight (from DFB 4) and the beat signal envelope can be used to obtainthe inverted version of the envelope at the continuous wavelength. Twooptical bandpass filters (OBPF 3 and OBPF 4) at the correspondingwavelengths can be used to separate the optical beat signal envelope andthe inverted optical beat signal envelope. Then, a temporal delay(Delay) of about 1/12 of the beat signal period (i.e., 8.3 ns for a 10MHz beat signal) can be introduced to the inverted optical beat signalenvelope before the two signals are equalized in amplitude and combinedat the optical coupler.

FIG. 11F shows the combined output after a photodetector (PD), with thebeat signal envelope 1218 and the amplitude rising/falling detectedoutput 1221. As shown, the rising amplitude has successfully resulted ina positive value (referred to as a “1”), while the falling amplitude hasresulted in a negative value (referred to as a “0”).

Logic Unit. In the phase unit 1106 and amplitude unit 1109 of FIG. 11B,the phase and amplitude information have been identified and encodedwith a “1” or “0” correspondingly, as summarized in the table of FIG.11G. The rest of the JAR system illustrated in FIG. 11B can be used to(i) determine the direction of frequency tuning based on the resultsfrom previous units, and (ii) enable or disable the JAR depending on howclose the jamming frequency is to the emitted frequency. Considering thetable of FIG. 11G, the relationship between amplitude, phase, and theneeded frequency change direction can be depicted as XOR logic. The XORlogic can be implemented using either electronics or photonics schemes;however, due to the low frequency nature of the output of the phase unit1106 and amplitude unit 1109 (e.g., with |f_(R)−f_(J)| mostly below 200MHz), electronic approaches are sufficient to implement the logic unit1112. The enabling of JAR can be governed by whether a jamming signal ispresent and if the jamming signal is within the jamming frequency range,f_(JAR). Once the frequency adjustment is in process, the logic unit1112 can also determine when to stop, e.g., once the jamming signal isno longer within the jamming frequency range of the new reference signalfrequency.

In the example of FIG. 11B, an Arduino Due board was used as the logicunit 1112. The logic unit 1112 can supply a control voltage (V_(VCO))for driving the VCO that generates the desired reference signalfrequency. Initially, the logic unit 1112 is set to output the V_(VCO)such that the VCO generates the reference signal 1115 at the initialfrequency. The JAR is enabled/disabled through the V_(ENABLE) input,which receives the output from the low pass filter (LPF) following thebeat signal envelope detector (ED). The bandwidth of the low pass filter(LPF) can be selected to match the desired jamming frequency rangef_(JAR). The actual f_(JAR) can be slightly adjusted with the use of anattenuator. If |f_(R)−f_(J)|<f_(JAR), then the beat signal envelopepasses through the LPF and serves as the enable input V_(ENABLE) to thelogic unit 1112.

When V_(ENABLE) is enabled, the logic unit 1112 takes the values at theV_(PHASE) and V_(AMPLITUDE) inputs that are connected to the outputs ofthe phase unit 1106 and amplitude unit 1109, respectively. The logicunit performs an XOR logic operation on the V_(PHASE) and V_(AMPLITUDE)inputs, and returns the desired frequency shift direction (e.g., “1”represents an increase in frequency, while “0” represents a decrease infrequency). The frequency incremental step can be set to be 1 MHz, whichcan be controlled by the V_(VCO) (e.g., at the Arduino Due board).Therefore, the V_(VCO) can keep increasing and/or decreasing (e.g., at astep of 1 MHz), updating the VCO driving voltage until|f_(R)−f_(J)|>f_(JAR), at which point the V_(ENABLE) will be disabled.

There are two scenarios in which V_(ENABLE) can be disabled: (i) Nojamming signal is present—the resultant “beat signal” will be solely thereference signal 1115 and the frequency will be too high to pass throughthe low pass filter (LPF); and (ii) If |f_(R)−f_(J)|>f_(JAR), thejamming signal is spectrally far from the reference signal 1115, and thebeat signal envelope will not pass through the low pass filter (LPF),which essentially disables the JAR via the V_(ENABLE) input. In eithercase, the logic unit 1112 will ignore the inputs at V_(PHASE) andV_(AMPLITUDE), and keep the latest V_(VCO) without further change to theVCO frequency.

Results of Photonic JAR System

The jamming avoidance capability of the photonic JAR system of FIGS. 11Aand 11B was tested using various types of jamming signals, including apure sinusoidal wave, a sinusoidal-amplitude modulated wave (with acarrier at 900 MHz and an amplitude modulation at 10 MHz), and adigitally-amplitude modulated signal (with a carrier at 900 MHz and anamplitude modulation at 10 Mb/s). FIG. 11H shows screen shots of the JARprocessing when the jamming signal is a pure sinusoidal wave. As shownin screen shots (i), (ii) and (iii), the jamming signal (f_(J)) firstapproaches from the lower frequency side (left) and moves towards thereference signal frequency (f_(R)) as illustrated in screen shot (i).The photonic JAR is enabled once |f_(R)−f_(J)| is smaller than f_(JAR),which pushes the reference signal to a higher frequency as illustratedin screen shot (ii). When the jamming signal (f_(J)) approaches from thehigher frequency side (right) and moves towards the reference signalfrequency, the JAR system pushes the reference signal (f_(R)) towards alower frequency as illustrated in screen shot (iii). Similar results areobserved in FIG. 11I when a sinusoidal-amplitude modulated signal isused as the jamming signal (f_(J)) and in FIG. 11J when adigitally-amplitude modulated signal was used as the jamming signal(f_(J)). The results show that the photonic JAR system successfullyenabled automatic adjustment of the reference signal frequency to avoidjamming.

Furthermore, a software-defined radio (SDR) was used to record aspectral waterfall of the photonic JAR system. FIGS. 11K-11M show thespectral evolution of the reference signal and jamming signal when thephotonic JAR was being triggered or not, with the vertical axisrepresenting the time evolution. As shown in screen shot (i) of FIG.11K, the reference signal (f_(R)) is at 1.205 GHz while the jammingsignal (f_(J)) is approaching the reference signal from the lowfrequency side. Initially f_(J) approaches f_(R), however the photonicJAR system is not triggered until f_(J) is higher than 1.056 GHz—whichis within f_(JAR). As f_(J) continues to increase, the photonic JARsystem automatically adjusts f_(R) to a higher frequency to keep f_(J)at least f_(JAR) away from f_(R). The adjustment stops once f_(J) stopsapproaching f_(R) and is at least f_(JAR) away.

For the scenario shown in screen shot (ii) of FIG. 11K, where thejamming signal f_(J) first approaches f_(R) from the lower frequencyside, stays at that frequency for a short while, and then moves awayfrom f_(R), the photonic JAR system will be triggered when f_(J) iscloser than 150 MHz and will move f_(R) away from f_(J). Once thedesired frequency separation is reached, f_(R) will stay at the newfrequency and will not be affected by f_(J) when it moves away fromf_(R). The photonic JAR system works similarly when the jamming signalis approaching form the higher frequency side as shown in screen shots(iii) and (iv) of FIG. 11K. Since the instantaneous bandwidth of the SDRis only 20 MHz, the frequency tuning range is intentionally limited towithin 20 MHz for displaying the whole JAR process. Similar evaluationswere conducted and recorded using the SDR when various modulationformats are used in the jamming signal, as shown in FIGS. 11L (with asinusoidal-amplitude modulated signal) and 11M (a digitally-amplitudemodulated signal). As can be seen, the photonic JAR system behavedsimilarly as in the scenario in FIG. 11K with a pure sinusoidal jammingsignal, showing its effectiveness towards various types of jammingsignals.

The photonic JAR system of FIGS. 11A and 11B can detect whether thejamming (or interference) frequency is spectrally within the jammingspectral range and can intelligently move the reference (ortransmission) frequency away from the jamming frequency. In this way,the reference signal will not cross the jamming signal frequency toavoid serious jamming. The photonic JAR system of FIGS. 11A and 11Bcomprises four functional units: the ZeroX unit 1103, the phasedetection unit 1106, the amplitude unit 1109, and the logic unit 1112,which can be implemented using semiconductor optical amplifiers (SOAs)by utilizing various optical nonlinear phenomena. The implementedphotonic JAR system was capable of escaping from sinusoidal, amplitudemodulated, and digitally-amplitude modulated jamming signals. Due to thewideband operation capability of photonics, the photonic JAR system canwork well with jamming signals in the MHz range to the tens of GHzrange.

It should be emphasized that the above-described embodiments of thepresent disclosure are merely possible examples of implementations setforth for a clear understanding of the principles of the disclosure.Many variations and modifications may be made to the above-describedembodiment(s) without departing substantially from the spirit andprinciples of the disclosure. All such modifications and variations areintended to be included herein within the scope of this disclosure andprotected by the following claims.

It should be noted that ratios, concentrations, amounts, and othernumerical data may be expressed herein in a range format. It is to beunderstood that such a range format is used for convenience and brevity,and thus, should be interpreted in a flexible manner to include not onlythe numerical values explicitly recited as the limits of the range, butalso to include all the individual numerical values or sub-rangesencompassed within that range as if each numerical value and sub-rangeis explicitly recited. To illustrate, a concentration range of “about0.1% to about 5%” should be interpreted to include not only theexplicitly recited concentration of about 0.1 wt % to about 5 wt %, butalso include individual concentrations (e.g., 1%, 2%, 3%, and 4%) andthe sub-ranges (e.g., 0.5%, 1.1%, 2.2%, 3.3%, and 4.4%) within theindicated range. The term “about” can include traditional roundingaccording to significant figures of numerical values. In addition, thephrase “about ‘x’ to ‘y’” includes “about ‘x’ to about ‘y’”.

Therefore, at least the following is claimed:
 1. A jamming avoidanceresponse (JAR) system, comprising: circuitry configured to transmit areference signal and receive an interference signal; photonic circuitryconfigured to generate optical spikes corresponding to positive zerocrossing points of the reference signal; and photonic circuitryconfigured to provide a phase output that indicates whether a beatsignal associated with the interference signal and the reference signalis leading or lagging the reference signal, the phase output based atleast in part upon the optical spikes and the beat signal.
 2. The JARsystem of claim 1, further comprising signal beating circuitryconfigured to generate the beat signal from the interference signal andthe reference signal.
 3. The JAR system of claim 1, further comprisingphotonic circuitry configured to generate optical pulses correspondingto rising edges of an envelope of the beat signal.
 4. The JAR system ofclaim 3, further comprising logic unit circuitry configured to determinean adjustment to a reference frequency of the reference signal based atleast in part upon the optical pulses and the phase output.
 5. The JARsystem of claim 3, wherein the photonic circuitry configured to generatethe optical pulses corresponding to the rising edges of the envelope ofthe beat signal comprises: a first electro-optic intensity modulator(EOM) configured to generate an amplitude modulated optical signal basedupon an envelope signal corresponding to the beat signal; a second EOMconfigured to generate an inverted amplitude modulated optical signalbased upon the envelope signal; and a non-linear optical device thatgenerates the optical pulses from a combined optical signal producedfrom the amplitude modulated optical signal and a delayed version of theinverted amplitude modulated optical signal.
 6. The JAR system of claim5, wherein the non-linear optical device is a length of highly nonlinearfiber (HNLF), a semiconductor optical amplifier (SOA), a silicon wirewaveguide or a photonic crystal fiber.
 7. The JAR system of claim 5,wherein the photonic circuitry configured to generate the optical pulsescorresponding to the rising edges of the envelope of the beat signalfurther comprises a bandpass filter configured to filter the opticalpulses from the non-linear optical device.
 8. The JAR system of claim 1,wherein the photonic circuitry configured to generate the optical spikescorresponding to the positive zero crossing points of the referencesignal comprises: an electro-optic intensity modulator (EOM) configuredto generate an amplitude modulated optical signal based upon thereference signal; a thresholder configured to clamp an amplitude of theamplitude modulated optical signal to generate a clamped optical signal;and a non-linear optical device that generates the optical spikes fromthe clamped optical signal.
 9. The JAR system of claim 8, wherein thephotonic circuitry configured to generate the optical spikescorresponding to the positive zero crossing points of the referencesignal further comprises a bandpass filter configured to filter theoptical spikes from the non-linear optical device.
 10. The JAR system ofclaim 1, wherein the photonic circuitry configured to provide the phaseoutput that indicates whether the beat signal is leading or lagging thereference signal comprises: an electro-optic intensity modulator (EOM)configured to generate an amplitude modulated optical signal based uponthe beat signal; and a semiconductor optical amplifier (SOA) configuredto generate the phase output from a combined optical signal producedfrom the amplitude modulated optical signal and the optical spikes fromthe photonic circuitry configured to generate the optical spikescorresponding to the positive zero crossing points of the referencesignal.
 11. The JAR system of claim 10, wherein the photonic circuitryconfigured to provide the phase output that indicates whether the beatsignal is leading or lagging the reference signal further comprises abandpass filter configured to filter the phase output from the SOA. 12.A jamming avoidance response (JAR) method, comprising: generatingoptical pulses that correspond to a rising edge of an envelope of a beatsignal associated with a transmitted reference signal and a detectedinterference signal; generating optical spikes that correspond topositive zero crossing points of the transmitted reference signal; andproviding a phase output that indicates whether the beat signal isleading or lagging the transmitted reference signal, the phase outputbased at least in part upon the optical spikes.
 13. The JAR method ofclaim 12, wherein generating the optical pulses comprises: generating anamplitude modulated optical signal based upon an envelope signalcorresponding to the beat signal; generating an inverted amplitudemodulated optical signal based upon the envelope signal; and generatingthe optical pulses from a combined optical signal produced from theamplitude modulated optical signal and a delayed version of the invertedamplitude modulated optical signal.
 14. The JAR method of claim 13,wherein a non-linear optical device generates the optical pulses fromthe combined optical signal.
 15. The JAR method of claim 14, wherein thenon-linear optical device is a length of highly nonlinear fiber (HNLF),a semiconductor optical amplifier (SOA), a silicon wire waveguide or aphotonic crystal fiber.
 16. The JAR method of claim 12, whereingenerating the optical spikes comprises: generating an amplitudemodulated optical signal based upon the transmitted reference signal;and clamping an amplitude of the amplitude modulated optical signal togenerate a clamped optical signal for generation of the optical spikes.17. The JAR method of claim 16, wherein a non-linear optical devicegenerates the optical spikes from the clamped optical signal.
 18. TheJAR method of claim 12, wherein providing the phase output comprises:generating an amplitude modulated optical signal based upon the beatsignal; and generating the phase output from a combined optical signalproduced from the amplitude modulated optical signal and the opticalspikes.
 19. The JAR method of claim 18, wherein the phase outputcomprises an optical spike having an amplitude that indicates whetherthe beat signal is leading or lagging the transmitted reference signal.20. The JAR method of claim 12, further comprising determining anadjustment to a reference frequency of the transmitted reference signalbased at least in part upon the optical pulses and the phase output,wherein the reference frequency is increased when the phase outputindicates a leading beat signal and the optical pulses indicate anincreasing amplitude, or when the phase output indicates a lagging beatsignal and the optical pulses indicate a decreasing amplitude.